Device for activating inductive loop sensor of a traffic light control system

ABSTRACT

A system and method for activating vehicle sensors which are based on inductive loops. A voltage controlled oscillator is scanned throughout the required frequency range while a balanced mixer, low-pass filter, and detector are arranged to show the presence of a signal from an inductive loop by the amplitude of the resultant signal at a difference frequency. The voltage controlled oscillator is then made to dwell at or near the frequency of the sensor loop by adjusting so as to keep the amplitude of that low-frequency difference as high as possible.

FIELD OF THE INVENTION

This invention relates to the field of automated traffic controlsystems, particularly those with vehicle presence sensors based oninductive loops buried beneath a roadway. More specifically, thisinvention relates to the field of devices for enhancing the ability ofsmaller vehicles to activate such sensors.

DESCRIPTION OF RELATED ART

Control of traffic signal lights by automatic vehicular sensors hasbecome very common throughout the world, as an alternative to signalcontrol based purely on timing.

Although many modern vehicle sensing systems are based on digitalcameras with pattern recognition software, many older and less costlysystems are based on detection of a change in inductance of a loop ofwire buried beneath the road surface. Various such sensors have beenavailable commercially. Some of the more widely-used models are the C800from 3M/Canoga, and the LM302 from Eberle Design, Inc. The operation ofthese sensors is described, for example, in U.S. Pat. No. 3,943,339(Koerner, Bienhoff, henderson, Higbee, and Koerner), U.S. Pat. No.3,984,764 (Koerner), and U.S. Pat. No. 3,989,932 (Koerner).

The basic operation of these systems relies on an inductive loop or coilof wire buried beneath the road surface, typically in a rectangularpattern with sides several feet long. This loop has some nominalinductance which will be reduced when some large metal object, such as avehicle, is brought into proximity, according to well known laws ofelectromagnetism. The loop is typically placed in an oscillator circuitsuch that its inductance directly affects the frequency of oscillation.The sensing circuitry then monitors the oscillation frequency andindicates presence of a vehicle when some sufficiently large and abruptchange takes place in that frequency. This is accomplished by eithermeasuring the time taken to accumulate some fixed number ofoscillations, or by measuring the number of oscillations in some fixedtime. Alternately, inductance change can be measured by applying a fixedfrequency signal, and monitoring a phase shift or bridge imbalance as iscommonly done in laboratory instruments for inductance measurement. Theimportant common feature is that some alternating current signal,typically in the 20 kHz to 200 kHz frequency range, is applied to theloop, and that the detection is triggered by some measurable change inthe frequency or phase of that current.

These systems are still very common, and can be expected to persist inmore emote areas where the cost of the visual systems may not bejustified. It is these less-traveled sites which present a particularproblem for the operators of smaller vehicles such as bicycles andmotorcycles.

Often a side-street or left-turn lane will have such a loop-basedsensor. Quite often, the sensitivity of these sensors will be set suchthat only vehicles as large as an automobile will cause sufficientperturbation to the magnetic field of the inductive loop to activate thedetector and cause the desired traffic light change. Often, thissensitivity is deliberately reduced to avoid maintenance problemsassociated with false triggering from large temperature changes oraging. A smaller vehicle, such as a bicycle, often will be unable toactivate the sensor, or the operator must resort to such inconveniencesas laying the bicycle down in the area of the loop with the hope thatsufficient inductance change will result. When the main street has alarge amount of traffic, and the side street or turn lane does not, thesituation can be very frustrating.

Various approaches have been taken to solving this problem. Frasier(U.S. Pat. No. 5,652,577) describes a passive device consisting of arelatively large metallic mesh or foil which is to be laid over the loopso as to simulate the presence of a large metallic object. This has themajor drawback of being very awkward and inconvenient.

Strang and Frus (U.S. Pat. No. 5,057,831) describes an active systemconsisting of two coils connected via an amplifier. One coil is used tosense the emitted alternating field from the loop, while the other isused to generate an amplified magnetic field which is then presented tothe loop by magnetic induction. Thus, the system tries to create theeffect of a large metallic object, which would reduce or exclude anyapplied field, by driving a somewhat canceling alternating field back tothe loop. This system has the advantage of being active and thus smallerthan the corresponding large metal object, but has the disadvantage ofbeing limited in the sensitivity which can be provided via the gain ofthe amplifier before self-oscillation takes place. In order to providemore sensitivity, greater separation between the two coils would benecessary, and the device would become larger. The adverse effects offeedback between output and input are discussed in '831 and described asregeneration. Although not based on regeneration, the inventors of '831describe it as difficult to avoid. It will be especially difficult toavoid regeneration, even with specific cancellation circuitry, across awide range of frequencies. Furthermore, relative orientation of the twocoils with respect to the loop is critical in order to ensure fieldcancellation.

Baer and Sunda (U.S. Pat. No. 6,072,408) describe two embodiments ofactive systems. In the first embodiment, a voltage controlled oscillator(VCO) is driven throughout the range of expected frequencies (20 kHz-200kHz) and connected to a transmitter and coil intent that, for some briefinstant, the frequency of the loop sensor will coincide with that of theVCO. At this point, the loop sensor circuitry will respond to thepresence of an inductively coupled signal much as it would to an abruptchange in inductance. If the loop sensor circuitry is sufficientlyunsophisticated, a brief perturbation in its current caused by couplingat or near its driven frequency would activate the detection circuitry.

In the second embodiment disclosed in '408, the VCO is controlled bycircuitry which temporarily or perhaps periodically shuts down thetransmitter and attempts to measure the actual frequency emitted by theloop sensor. With this measurement, the transmitter is then set to asimilar frequency so that it can dwell on or near the correct frequencyso as to further enhance activation. These embodiments both suffer fromdifficulties associated with some of the more advanced aspects of loopsensor technology as described below.

In most modern implementations, loop sensor technology implements atime-sharing approach to vehicle presence detection. There are severalreasons for this, but the primary reason is that a single intersectionmight have many detection zones. For example, there may be two left-turnlanes with need to distinguish between vehicle presence in each. Bytime-sharing the loop-sensing function, interaction between two adjacentloop sensing circuits can be avoided. Each loop is periodicallyactivated for a brief time “window” (typically 5 ms to 50 ms), followedby a large interval of inactivity during which time other loops can beactivated. During that activation window, the loop receives a brief“burst” of high-frequency signal and detection determination is madeduring that burst time. Adjacent loops will be activated in sequence, sothat never will two be activated at the same time, and the possibilityof inadvertent interaction and false-triggering is eliminated.

Additionally, many installations will have timing requirements such thata vehicle must be detected for a significant duration of time (a dwelltime) before activation occurs. This is to ensure that a signal lightchange is required, and that the vehicle didn't just briefly trigger thedetector and then make an unusual move (such as a last-minute lanechange or, worse, a red signal transgression). Furthermore, someintersections require that a first loop sensor be activated before avehicle dwells on a second sensor as if to ensure that the vehicleactually approached from the approved direction.

For these reasons, the devices of Baer and Sunda would havedifficulties. The first embodiment would not provide a sufficientduration of detection for those systems requiring a dwell time. Thesecond embodiment would have difficulty with the large number of systemswhich employ time-sharing. In order to capture the frequency of the loopsensor oscillator, their embodiment sampled an input from an inductivelycoupled coil for some time interval and counted the cycles-received.With a time-shared signal, which is not continuous but only presentduring some low duty cycle, the time interval of the measurement must besomehow matched to that of the loop sensor burst so that theinstantaneous frequency of the burst can be measured. This wouldundoubtedly require measurement circuitry far beyond what is disclosedin the '408 patent, especially since burst durations vary from system tosystem.

SUMMARY OF THE INVENTION

The present invention accomplishes loop sensor activation by generatingan out-put signal of large amplitude and applying it to a coilpositioned for inductive coupling to the sensor loop. The frequency ofthe output signal is brought close to that of the loop sensor byadjusting a VCO while simultaneously monitoring the difference frequencysignal derived by heterodyne mixing of the generated signal with thesignal from the loop sensor. This difference frequency signal isprocessed through a low-pass filter designed to only allow signals ofsufficiently low frequency to be amplified. When the frequency of thismixer product is within the passband of the low-pass filter, itsamplitude is measured in a detection process, and averaged over a timeperiod corresponding to many cycles of the time sharing described above.This measurement is performed by a small controller, such as amicrocontroller, configured to also correct the frequency of the VCO soas to maximize the amplitude of the signal at the difference frequency.In this manner, a frequency very close to that of the loop sensor isfound, and a signal at that frequency is simultaneously retransmitted tothe loop sensor over an arbitrarily long dwell time. By proper choice ofthe corner frequency of the low-pass filter, the difference between theloop sensor frequency and resultant VCO frequency can be made smallenough for activation of the vehicle presence detection circuitry tooccur.

In contrast with prior art activation techniques, the present inventionsimultaneously transmits while monitoring its frequency relative to thatof the loop sensor. There is no need for a pause in operation to measurethe loop sensor frequency. Furthermore, by the use of a balanced mixingarrangement, a single coupling coil can be used and there is no need fora second inductive coupling coil for communication with the loop sensor.

A further major benefit of the present invention is that the sensitivityis not limited by stability concerns. By using the heterodyne mixingprinciple, gain is applied only to signals at the difference frequency.This difference frequency, when present, is far below the operatingfrequency, and so there is no need to keep the outgoing signal separatedfrom the incoming signal, aside from the amount of isolation requiredfor proper operation of the mixer. As a result, this gain can be quitehigh, as can the sensitivity, while the output amplitude is kept at itsmaximum value throughout the operating cycle, regardless of distancefrom the sensor loop.

Furthermore, there is no need for a circuit according to the presentinvention to calibrate its VCO in order for an input frequencymeasurement to be mapped to an appropriate VCO setting. By virtue of theheterodyne mixing and low-pass filtering, the VCO frequency which leadsto significant amplitude of a signal at the difference frequency isautomatically very close to the input frequency.

A further benefit of the present invention is that time-shared signalswith even modest duty cycles are easily accommodated by ensuring thatthe time constant of the amplitude detection circuitry is sufficientlylarge. This serves to average out the effects of the short bursts insuch a manner that the actual timing of the bursts is irrelevant.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a block diagram of a basic activation device according tothe invention, showing inductive coupling coil 1, inductor 2, VCO 3,balanced drive circuitry 4, balanced mixer 5, low-pass filter 6,amplifier 7, detector 8, and microcontroller 9.

FIG. 2 shows a representative time-shared loop sensor signal 10,corresponding beat frequency signal 11, and detector output 12.

FIG. 3 shows details of an exemplary embodiment of the invention.

FIG. 4 shows a flowchart of an exemplary embodiment of the algorithmused by microcontroller 9.

FIG. 5 shows a block diagram of an alternative embodiment of theinvention, with automatic activation circuitry provided by logic gates47 and 48, master output control signal 1101, analog switch 1102, highfrequency amplifier 1103, and detector 1104.

FIG. 6 shows details of an exemplary embodiment of the inventionincluding signal 96 from microcontroller 9 which enables or disables thebalanced drive circuitry 4 for power conservation purposes.

FIG. 7 shows a flowchart of an exemplary embodiment of the algorithmused by microcontroller 9 when power conservation is used with signal96.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 shows a block diagram of an exemplary embodiment of theinvention. Coil 1 serves to both carry the high-power output signalbeing sent to the loop, as well as to receive the much weaker signalemitted by the sensor loop. Balanced drive circuitry 4 takes the signalfrom VCO 3, which is constrained to have a frequency covering all theexpected operating frequencies of loop sensors (typically 20 kHz-200kHz) and applies it to both coil 1 and inductor 2. Coil 1 consists ofapproximately 20 turns of wire wound in a rectangular form,approximately 4 inches by 5 inches along its sides. The nominalinductance of coil 1 is chosen to be approximately 50 microhenries.Inductor 2 is wound on an iron-powder toroidal form and also has aninductance of approximately 50 microhenries. The field of inductor 2should be kept substantially isolated from coil 1 so that little mutualinductance exists between the two. In this manner, substantiallyout-of-phase signals applied to coil 1 and inductor 2, as connected inseries, will produce little signal at the output frequency at theircommon terminal.

In practice, some imbalance will occur, and the common connection pointwill contain some amount of the output signal. However, this will notadversely affect the operation of the circuit as long as it issufficiently low as to keep mixer 5 from becoming desensitized.

Meanwhile, coil 1, when placed in proximity to an active sensor loop,will also develop some amount of signal at the frequency of the sensorloop at the common terminal with inductor 2 and, hence, at the firstinput to mixer 5. The second input to mixer 5 comes directly from VCO 3and hence carries a large signal with a frequency corresponding to theoutput frequency. Mixer 5 then mixes these two signals in a heterodyneprocess to produce both sum and difference frequencies at its output,according to methods well known in the design of radio receivers (See,for example, Carlson, A. B. “Communication Systems”, pp. 201-202,McGraw-Hill 1975).

Low-pass filter 6 then substantially attenuates all but the differencefrequency signal. When that difference frequency is within the passbandof low-pass filter 6, namely less than approximately 2 kHz, it isamplified by amplifier 7 and presented to detector 8. Detector 8 can bemuch like the detector found in inexpensive AM radio receivers, commonlyknown as a peak detector. Its output is a slowly varying representationof the amplitude of its input. When the low frequency alternatingsignal, derived from the frequency difference, is presented to detector8, its output voltage rises. That is, the difference frequency of a fewhundred to a few thousand Hertz at the input to detector 8 causes itsoutput to quickly rise to some positive voltage, which decays relativelyslowly after the input signal is removed. As a result, even lowduty-cycle time-shared bursts of high frequency signal presented to coil1 will result in some rise in voltage at the output of detector 8 which,although unsteady, will remain above zero for a time substantiallylonger than the cycle time of the time-sharing sequence.

Output from detector 8 is presented to microcontroller 9. Ideally,micro-controller 9 contains an analog-to-digital converter which can,under software control, measure the output voltage of detector 8.Microcontroller 9 also contains a digital-to-analog converter (DAC)which is connected to VCO 3.

The software executed by microcontroller 9 is designed to slowly scanthe output of its DAC through the range of voltages corresponding to therange of frequencies (typically 20 kHz-200 kHz) expected for loop sensoroperation, according to the operation of VCO 3. When some, albeit small,indication of a beat frequency signal is indicated by detector 8, thesoftware then immediately stops the scanning, and implements any ofvarious algorithms for fine-tuning the output frequency to closely matchor dither around that of the input. In this manner, the frequency orphase measurement circuitry in the loop sensor is presented with anabrupt perturbation and if sufficiently strong, a sensor activation willresult.

FIG. 2 shows three signals corresponding to the acquisition of an inputsignal. Shown at 10 is a periodic burst of high frequency signal, aswould be derived from coil 1 in the presence of a time-shared sensorloop. At 11 is shown the beat signal, as filtered by low-pass filter 6and amplified by amplifier 7. This signal consists of a relatively smallnumber of cycles of the difference frequency corresponding to thedifference between the input signal from a loop sensor and theinstantaneous frequency of VCO 3. At 12 is shown the output of detector8 corresponding to signal 11. A signal, with a net positive value, isbuilt up quickly and decays over a time somewhat longer than theinter-arrival time of input bursts.

FIG. 3 shows details of a preferred embodiment. VCO 3 is implemented asa CMOS integrated circuit (IC) 31, such as the MM74HC4046M fromFairchild Semiconductor Corporation. Its control voltage input isderived from the filter consisting of resistor 92 and capacitor 93.Microcontroller IC 91 can be, for example, a PIC16F716 from MicrochipCorporation, which has a pulse-width modulation controller internallyconfigured to operate as a low-cost digital-to-analog converter (DAC) bythe use of resistor 92 and capacitor 93. VCO IC 31 is configured withresistors 32 and capacitor 33 so that its frequency varies from 20 kHzto 200 kHz when its input control voltage is driven over the entirerange of voltages from this DAC output. In this manner, microcontrollerIC 91 can control the frequency of VCO 3 over the full range of requiredoperating frequencies.

The output of VCO 3 is connected to balanced drive circuit 4 comprisinglogic gates 45 and 46, MOSFET driver transistors 41, 42, 43, and 44, andtiming components in the form of resistors 47 and capacitors 48. LogicNAND gates 46, for example the SN74HC00D from Texas InstrumentsCorporation, form a non-overlapping complementary drive circuit inconjunction with resistors 47 and capacitors 48. In this configuration,each output substantially represents the complement of the other, exceptfor a brief time, typically several hundred nanoseconds, around thetransition time. During this interval, both outputs are held high, thusensuring that there is a small window of time between one output goinghigh and the other going low. These outputs then drive p-channelenhancement MOSFETs 41 and 43, and, through the use of inverters 45 (forexample the SN74HC04D from Texas Instruments Corporation), n-channelenhancement MOSFETs 42 and 44.

P-channel enhancement MOSFETs 41 and 43 are, for example, of typeNDS332P from Fairchild Semiconductor Corporation, while N-channelenhancement MOSFETs 42 and 44 are, for example, of type NDS331N alsofrom Fairchild Semiconductor Corporation. Both of these devices arecapable of very large drain currents, as each has a typicalon-resistance of less than 1Ω. As a result, it is important that eachcomplementary pair, 41 and 42 or 43 and 44, not be driven in such amanner that both devices of the pair are turned on at the same time.This is the reason for the non-overlapping drive circuit outlined above.It prevents unnecessary current flow, potentially draining an operatingbattery or damaging the devices.

Transistors 41, 42, 43, and 44 form what is commonly referred to as anH-bridge. Each common connection of a complementary pair issubstantially opposite in phase to the other. Thus, aside from the briefnon-overlap time described above, when the connection between MOSFETs 41and 42 is low, that between 43 and 44 is high, and vice-versa. A verylarge amount of current, however, can be delivered to whatever load isconnected between the pairs. As they are driven from logic gates 45 and46 and, in turn, from IC 31, the voltage across the H-bridge will be aroughly square wave signal with a frequency corresponding to that of IC31.

Across the output of this H-bridge is placed coil 1 and inductor 2.Inductor 2 can be a toroidal inductor of the type used in switchingpower supplies, such as the model 2209-H from J. W. Miller Magneticsdivision of Bell Industries. By having similar nominal inductances, asdescribed above, the voltage at the common connection between coil 1 andinductor 2 exhibits relatively little of the drive signal.

Operational amplifiers 52 and 53, such as the MCP6284 from MicrochipCorporation, buffer the signal at the common connection of coil 1 andinductor 2, and form its inversion. Multiplexer 51 can be, for example,the SN74LVC1G3157DBV from Texas Instruments, Corporation. The inputs tomultiplexer 51 are thus 180-degrees out of phase with respect to eachother and allow multiplexer 51 to act as a balanced mixer. The controlsignal for multiplexer 51 is taken from the output of IC 31 and, hence,VCO 3. Any residual signal at the frequency of VCO 3 present at theinput to multiplexer 51 will then be synchronously converted to D.C. andremoved afterward by suitable coupling. By this action, the residualoutput signal present at the common connection of coil 1 and inductor 2is prevented from being applied to the following amplificationcircuitry.

However, any other signal generated by coil 1, as derived from mutualcoupling to a sensor loop, will result in a mixer product at the outputof multiplexer 51. This product will contain the difference frequencybetween that of VCO 3 and that of the sensor loop. This signal is thenpresented to low-pass filter 6.

Low-pass filter 6 is comprised of operational amplifier 61, possibly ofthe same type as 52 and 53 (although its frequency response need not benearly as high as those), and various resistors and capacitors to makean active low-pass filter with an upper cutoff frequency of a severalkHz. The output coupling, via capacitor 69, and bias circuitry, viaresistors 66 and 67 and capacitor 68, are such that there is also alow-frequency cutoff frequency chosen to be a few hundred Hz or below.This is what serves to remove the synchronous mixer product describedabove, but is easily chosen to be so low as to not adversely affect theoperation of the circuit.

The output of low-pass filter 6 is connected to amplifier 7, comprisingoperational amplifier 71, possibly of the same type as 61, resistors 72and 73 which determine its gain, and stabilizing capacitor 74. Theoutput of this stage is then a highly filtered and amplified version ofthe difference frequency signal as generated by the output of VCO 3 andwhatever input signal might be present. Signal 11 of FIG. 2 depicts thisoutput in the presence of a pulsing input signal with frequencysufficiently near to that of VCO 3.

Detector 8 utilizes two Schottky diodes, such as the type BAT54 fromVishay Semiconductor Corporation. Capacitor 84 and resistor 85 serve toset the decay rate of this detector, and is set to provide decay oversome tens of milliseconds. Output from detector 8 goes to theanalog-to-digital input of microcontroller IC 91.

FIG. 4 shows a flowchart of a typical program to be executed bymicrocontroller IC 91. At power-up, initialization of all control bitsand timing registers takes place, and then the program enters into asoftware loop which consists of a short wait of several milliseconds, atest of its input for presence of signal, and barring the detection ofone, a step to the next frequency, either increasing or decreasing. Inthis manner, the microcontroller effectively scans through all relevantfrequencies, in an increasing and then decreasing direction, until somesignal is presented to coil 1 which generates an output from detector 8which is above some pre-determined threshold.

Upon detection of some event, indicating a nearby sensor loop, anothersoftware loop is entered which attempts to dwell at or near thefrequency of the discovered signal, and to possibly improve thefrequency of VCO 3 to more closely approach that of the input. As shownin FIG. 4, when a signal is detected, control flows to the right andenters a second test to ensure that the signal remains present. Assumingit does, three frequency values are computed. The first is the DAC valuecorresponding to the current frequency where signal was discovered. Thesecond is one increment below that of the current frequency, and thethird is one increment above.

The loop proceeds by first dwelling at the current frequency, and takinga long average of the amplitude of the output of detector 8, with manysamples over a period greater than 100 ms. VCO 3 is then set to thesecond frequency, still close to the original, and a similar longaverage reading of amplitude is taken. If this amplitude is greater thanthat at the first frequency, the second frequency becomes the currentfrequency, and the loop begins again.

If the amplitude as measured for the first frequency is greater thanthat of the second, then the third frequency is tried. If itsmeasurement is greater than that of the first, the third frequencybecomes the new current frequency, and the loop begins again. If not,then the original frequency is retained and the loop begins again.

If, during the test at the beginning of this second software loop, theinput signal disappears, the current frequency is retained for someamount of time (for example, 200 ms), and control is returned to theoriginal software loop. In this manner, a brief lapse in signal, such asis encountered when moving between a first and second loop sensor asfound in some intersections, will not necessarily lead to a completelynew search, with its associated delays, and possible loss of detectionby the loop sensor circuitry.

As described, the software is designed to scan sufficiently slowly thata detected signal is not passed over by the time the second softwareloop is entered. That is, the response speed of the DAC and its outputfilter, as well as that of detector 8, must be factored in to thescanning speed of the first software loop. It is possible to implement avariation of this program in which scanning happens at a faster rate,but which would require a modification of the second software loop to goback a few steps to find the frequency at which the detection actuallyoccurred. Many such speed improvements can be implemented with moresophisticated software and/or hardware.

In practice, it is found that an important advantage is obtained byperforming the scanning primarily in the direction of decreasingfrequency. That is, the initial frequency at activation should be at thehigh end of the range of possible frequencies, and the scan steps shouldeach act to decrease the frequency while the system searches for outputfrom detector 8. Upon reaching minimum frequency, in the absence ofoutput from detector 8, the frequency is made to jump back to thehighest frequency and, after a short pause to allow any transientscaused by the large frequency jump to decay, step-by-step scanning inthe direction of decreasing frequency resumes. The advantage of scanningin the direction of decreasing frequency is due to the possibility ofhaving a false detection due to mixing of the incoming signal from theloop sensor with a harmonic of the VCO frequency. When mixer 5 isimplemented with such circuitry as multiplexer 51, the third harmonic ofthe output signal of VCO 3 can cause a substantial beat frequency signalto be generated when an input signal is present. However, no such falsedetection can be expected from the third harmonic of the loop sensorsignal mixing with the fundamental of VCO 3, since signals from loopsensors are generally sinusoidal and have very little harmonic content.Thus, by scanning primarily in the direction of decreasing frequency,the first detector output to be encountered will be most likely that forwhich the fundamental VCO frequency is close to that of the loop sensor.In this manner, the stronger fundamental VCO signal is available tointeract with the loop sensor and inadvertent operation with the weakerharmonics is substantially avoided.

Heretofore, initiation of the circuitry of the invention by the user wasunspecified. In cases where the device is powered from a small battery,it is important that the high current demand of the drive circuitry 4,as well as the remaining analog processing circuitry comprising mixer 5,low-pass filter 6, and amplifier 7, be reduced or eliminated when a loopsensor is not present. This functionality could be provided bypushbutton 1002 connected to battery 1101 as shown in FIG. 3. When theuser approaches a loop sensor, the user will press pushbutton 1002, andthe operation commences.

It is possible for automatic initiation to be provided. FIG. 5 shows ablock diagram of a system according to the invention which would sensethe presence of an inductive loop sensor, and activate the circuitrywhich consumes the largest share of operating current only subsequently.Microcontroller 9 would be arranged, by its own software, to enter a“sleep” mode wherein its own current consumption reduces to only a fewmicroamperes or less. Before doing so, it would de-assert power controlsignal 1101, which would force the outputs of logic gates 47 and 48 tobe such that all four power MOSFETs 41, 42, 43, and 44, were turned offregardless of the signals present on the other inputs of gates 47 and48. Similarly, power to most of the other circuitry, including VCO 3,mixer 5, low-pass filter 6, and amplifier 7 are controlled from signal1101 by the use of switching circuitry shown as 1106 in FIG. 5. Whensignal 1101 is de-asserted, power to VCO 3, mixer 5, low-pass filter 6,and amplifier 7, all powered from switched power signal 1107, isdisconnected from primary power source Vcc. Before going into “sleep”mode, microcontroller 9 would also, under software control, assertsignal 1105 thus enabling analog switch 1102. High frequency amplifier1103 must be designed to consume very little current while providingsufficient gain across the range of desired operating frequencies(typically 20 kHz to 200 kHz) that detector 1104 would produce alogic-level output when coil 1 is placed in position above an activesensing loop. This logic level output from detector 1104 is then used toactivate microcontroller 9 through an interrupt, removing it from its“sleep” mode. Upon leaving “sleep” mode, microcontroller 9, asdetermined by its software, would then assert power control signal 1101,de-assert signal 1105, and normal operation of the device according tothe algorithm described earlier would ensue. After a sufficiently longperiod of loss of signal from detector 8, while under normal operation,the software of microcontroller 9 would then cause it to de-assertsignal 1101, assert signal 1105, and put it back into “sleep” mode forpower savings.

A circuit diagram of an alternate technique for automatic initiation isshown in FIG. 6 in which circuitry is provided to allow microcontroller9 to selectively inhibit the coupling between VCO 3 and coil 1. In thisembodiment, logic gates 45 have been relocated, and logic gates 49 havebeen added to balanced drive circuitry 4. In the arrangement shown,signal 96 from microcontroller 9 is asserted to enable the signal fromVCO 3 to be applied to MOSFET driver transistors 41, 42, 43, and 44. Inthis state, operation is as earlier described in connection with FIG. 3.

When de-asserted, signal 96 then forces MOSFET driver transistors 41 and43 to be turned off, while MOSFET driver transistors 42 and 44 areturned on. As a result, the high power output signal is not carried bycoil 1 as long as signal 96 is de-asserted. However, VCO 3 remainsconnected to mixer 5, and signals induced in coil 1 continue tocontribute to the voltage at the input to mixer 5. A signal at the beatfrequency will still be generated at the output of amplifier 7, anddetector 8 will continue to indicate the presence of a signal from aloop sensor.

In the absence of a signal from a loop sensor, this embodiment willcontinue to search, much as in the embodiment of FIG. 3. However, thesoftware executed by microcontroller 9, as shown in flowchart form inFIG. 7, will cause microcontroller 9 to assert signal 96 in response toa sufficiently strong signal from detector 8. At that point, operationcontinues with balanced drive circuit 4 enabled, and large currents willflow in coil 1 at the frequency of VCO 3. When a signal no longer existsat the output of detector 8, signal 96 is de-asserted, and a short pausein operation is performed in order to prevent any transient signalscaused by the abrupt cessation of output from balanced drive circuit 4from regenerating an output from detector 8. After that pause,microcontroller 9 resumes scanning across relevant frequencies untilanother detection occurs.

Thus, in the absence of a loop sensor signal, no large current flows incoil 1, and power consumption is greatly reduced. With MOSFET drivers 41and 43 turned off, the remaining circuitry can be made to draw verylittle current. Pushbutton 1002 is replaced in this embodiment by switch1003 owing to the lack of need for the user to de-activate the devicefor power conservation purposes when not over a loop sensor.

De-activation of balanced drive circuit 4 as shown in the embodimentshown in FIG. 6 is implemented primarily for power conservationpurposes. The operation of mixer 5 and filter 6 ensures that detectioncan be performed whether balanced drive circuit 4 is activated or not.This is in contrast to the prior art of Baer and Sunda (U.S. Pat. No.6,072,408) in which a method for disabling that transmitter is describedfor purposes of measurement of the loop sensor frequency during a timinginterval. Thus, as described in '408, that transmitter is occasionallyinhibited while in the presence of a loop sensor in order to measure itsfrequency. In contrast, a system according to the present invention isable to determine the presence and frequency of the loop sensor signalwithout the necessity of inhibiting its own output signal. It should beunderstood that numerous changes in details of construction and thecombination and arrangement of elements and materials may be resorted towithout departing from the true spirit and scope of the invention ashereinafter claimed.

1. A system for activating a traffic presence sensor of the type basedon an inductive loop, the system comprising: an oscillator coupled to acoil; a mixer coupled to said oscillator and to said coil; a filtercoupled to said mixer; a detector coupled to said filter which detectsan output of said filter; and a controller which controls the frequencyof said oscillator based on measurements taken from an output of saiddetector.
 2. The system according to claim 1, wherein: said filtercomprises a low-pass filter.
 3. The system according to claim 1,wherein: said oscillator is a voltage controlled oscillator.
 4. Thesystem according to claim 1, wherein: said controller comprises amicrocontroller.
 5. The system according to claim 1, wherein: said mixeris a balanced mixer.
 6. The system according to claim 1, wherein: saidmixer and said filter are configured so as to generate a signal having afrequency equal to a difference between the frequency of said oscillatorand that of a nearby traffic presence sensor.
 7. The system according toclaim 6 wherein: said detector is configured to generate an output whichis related to the amplitude of said signal generated from said mixer andsaid filter.
 8. The system according to claim 7 wherein: said controllercomprises a microcontroller; and said microcontroller executes a programwhich continually adjusts the frequency of said oscillator so that saidoutput of said detector dwells at or near that value which indicatesmaximum possible amplitude of said signal.
 9. The system according toclaim 1 further comprising: power control circuitry for placing saidsystem into a state during which functionality is suspended and powerconsumption is greatly reduced; a detector for detecting the presence ofa traffic presence sensor; and interrupt circuitry for causing saidsystem to leave said state upon receiving a signal from said detectorfor detecting the presence of a traffic presence sensor.
 10. The systemaccording to claim 1 further comprising: an apparatus connected to saidcontroller for inhibiting coupling between said oscillator and saidcoil.
 11. A method for activating a traffic presence sensor of the typebased on an inductive loop comprising: generating a first signal withadjustable frequency; impressing said first signal onto a coil; bringingsaid coil near to the inductive loop of a presence sensor; combiningsaid first signal with output of said coil in such a manner as togenerate a second signal substantially containing the difference infrequency between said first signal with adjustable frequency and thefrequency of the current driving said inductive loop; measuring theamplitude of said second signal; and adjusting said frequency of saidfirst signal such that said difference in frequency is small enough tocause activation of said traffic presence sensor.
 12. A method accordingto claim 11 wherein: said step of combining further includes a step oflow-pass filtering.
 13. A method according to claim 11 wherein: saidstep of adjusting said frequency is accomplished by adjusting thevoltage of a voltage controlled oscillator.
 14. A method according toclaim 11 wherein: said steps of measuring and adjusting are accomplishedthrough the use of a microcontroller.
 15. A method according to claim 11wherein: said step of combining comprises a step of balanced-mixing. 16.A method according to claim 11 further comprising the step of: enablingsaid step of impressing only when said said step of measuring theamplitude of said second signal indicates the recent proximity of anactive traffic presence sensor.